MOS linear region impedance curvature correction

ABSTRACT

A system and method to correct or cancel MOS linear region impedance curvature employing an analog solution to trim out the MOS linear region impedance curvature while accommodating PVT spreads in values of internal or external precision resistors. The linear region curvature correction may be obtained by using two MOS transistors in the pad driver/buffer and operating the transistors so as to proportionately increase output impedance of one of them when the output impedance of the other decreases, and vice versa. A linear pad impedance may be maintained over a range of Vpad values, while also maintaining the Vgs supplied to pad driver transistors at its maximum possible value to obtain greater linearity. The approach of the present disclosure relaxes the requirements on the voltage/current references used in the MOS pad drivers and makes tight impedance control possible, especially in a situation where the MOS fabrication process (typically all currently used processes) does not have available an internal precision resistor with a reasonably well controlled value.

This application is a divisional of U.S. application Ser. No. 10/804,699filed Mar. 19, 2004, now U.S. Pat. No. 7,282,948 and entitled MOS LinearRegion Impedance Curvature Correction.

BACKGROUND

1. Field of the Disclosure

The present disclosure broadly relates to pad impedance in MOS (MetalOxide Semiconductor) driver circuits and, more particularly, to anapparatus and method to correct MOS linear region impedance curvature.

2. Brief Description of Related Art

The high speed I/O standards governing data transfers between two highspeed electronic devices require a tight control of output impedance ofa device driver or buffer circuit in such devices. In modern I/Ostandards, the output impedance is required to be controlled to anominal value ± some percentage error or variation. For example, FIG. 1illustrates an exemplary pad impedance specification 10 under USB2.0(Universal Serial Bus version 2.0) for a low impedance driver circuit.An example of a low impedance drive unit is a USB1.1 (Universal SerialBus version 1.1) buffer circuit within a USB2.0 PHY (Physical Layer)device. The specifications mandated in the chart in FIG. 1 may apply tosuch USB1.1 buffer circuits. In USB2.0, the output impedance of a drivercircuit or buffer unit must be controlled within 45 Ohms ± 10% oftolerance. Therefore, the allowable range of output impedance becomesfrom 40.5 Ohms to 49.5 Ohms. This allowable range is shown by the shadedportion in FIG. 1. It is observed from FIG. 1 that such impedancecontrol applies over a certain voltage range on a drive circuit outputpad (i.e., the voltage range of Vpad shown in FIG. 2 and discussed laterhereinbelow.) In case of USB2.0, this voltage range is from 0V to 1.1V.Beyond Vpad=1.1V, the pad impedance (Rpad in equation (1) givenhereinbelow) is allowed to increase as seen from the spread of theshaded region in FIG. 1.

The discussion herein uses the terms “pad” or “integrated circuit pad”interchangeably to refer to an electrically conducting junction oroutlet for a circuit that is fabricated using a semiconductorfabrication process (e.g., a CMOS fabrication process). The pad for thecircuit allows the circuit to be connected to another circuit on thesame integrated circuit (IC) chip or to another circuit or deviceexternal to the chip containing the circuit with the terminating pad.

FIG. 2 illustrates a general approach to pad impedance control in a MOSdriver circuit 12. The MOS driver circuit 12 is fabricated as part of anintegrated circuit (IC) and includes an n-channel MOS transistor M1 (14)having its gate terminal 16 connected to a bias voltage Vgs (denoted asa gate-to-source voltage source 17), its source terminal 18 held at aground potential, and its drain terminal 20 connected to the integratedcircuit's or chip's output pad (IC pad) 24 via a series resistor Rs22—also referred to interchangeably herein as an internal precisionresistor, an internal linearizing resistor or an internal terminationresistor. The internal linearizing resistor 22 is typically integrallyfabricated (i.e., on the same chip) with the MOS transistor 14. Asdiscussed hereinbelow, an external precision resistor (not shown) mayalso be used instead of the internal one. The voltage present at theoutput pad (e.g., when another electronic device is connected to the ICcontaining the driver circuit 12) is denoted by the variable pad voltagesource Vpad 26. In the circuit configuration of FIG. 2, the impedancepresent at the IC pad 24 may be given by the following equation:

$\begin{matrix}{R_{pad} = {\frac{V_{pad}}{I_{pad}} = {R_{s} + \frac{1}{\beta\left( {V_{gs} - V_{T} - \frac{V_{ds}}{2}} \right)}}}} & (1) \\{{{where}\mspace{14mu}\beta} = {\left( \frac{\mu_{o}}{C_{ox}} \right)\left( \frac{W}{L} \right)}} & (2)\end{matrix}$It is noted here that equation (1) neglects the effects of channellength modulation, λ. In equation (1), Rs is the internal linearizingresistor 22, Vgs is the gate-to-source voltage 17, V_(T) is theappropriate threshold voltage (for NMOS or PMOS), Vds is thedrain-to-source voltage present in the transistor M1, and β is thetransconductance of the MOS device 14 (in mA/Volt). As is known in theart, the value of β is as defined in equation (2), in which μ₀ is thesurface mobility of the channel (holes for p-channel or electrons forn-channel) given in (cm²/volt.second), C_(ox) is the MOS gatecapacitance per unit area (in F/cm²), W is MOS device's effectivechannel width (cm), and L is MOS device's effective channel length (cm).

It is noted that different MOS fabrication technologies may havedifferent values for μ₀ and C_(ox). If μ₀ and C_(ox) are low, then thewidth-to-length ratio W/L (i.e., the size of the MOS device) needs to bemade very large to compensate for the lower μo and Cox values, therebymaintaining the impedance control for Rpad in equation (1). It isfurther noted that the equations (1) and (2) apply to the MOS transistorM1 in its linear region (i.e., the region between cutoff and saturation)of operation. The linear region of operation of the MOS transistor 14with an internal linearizing resistor 22 is shown in the two pad or“output” current (Ipad) versus pad or output voltage (Vpad) graphs (thetransconductance graphs) 28, 30 in FIG. 2. In the linear region, thecurvature of the Ipad-Vpad graph 28 is small with large value of Vgs 17,whereas the curvature of the Ipad-Vpad graph 30 is large with lowervalues of Vgs 17. This curvature is interchangeably referred tohereinbelow as “MOS linear region impedance curvature,” or “MOS linearregion curvature,” “linear region curvature,” or simply “impedancecurvature.” It is known that the slope of an Ipad-Vpad curve gives thereciprocal of Rpad. Thus, because of the MOS linear region curvature,the values of Rpad are not the same or controlled for graphs 28 and 30.

Although the series resistor Rs 22 forces the NMOS transistor M1 to bein its linear region, for this technique of impedance (Rpad) control towork, it is usually assumed that Vgs is at its maximum value (e.g., atsupply voltage Vdd in an NMOS device) and that the PVT (Process VoltageTemperature) variation of Rs 22 is relatively small (at least withrespect to the PVT impedance variation of transistor M1). Thegate-to-source voltage Vgs 17 must normally be at its maximum possiblevalue (for transistor M1 linear region operation) to minimize theimpedance curvature of transistor M1 (as, for example, in the graph 28in FIG. 2) because of the changing Vds as the pad voltage Vpad 26changes (due to, for example, fluctuations in the operatingcharacteristics of another device or circuit stage connected to the MOSdriver circuit 12 through the IC pad 24 or presence of a “return wavefront” from such device operating as a signal receiving device).

The effect of PVT spread of the internal linearizing resistor Rs 22 maybe illustrated in the case of a USB1.1 buffer within a USB2.0 PHY asfollows. In a 0.11 μm embedded CMOS fabrication technology, the value ofRs 22 will spread by as much as 45% from its nominal value over PVT.Thus, for example, if the nominal value (at 0° C.) of Rs=28 Ohms, thenit will actually PT spread between about 22 Ohms and 35 Ohms. Theproblem is the lower 22 Ohm limit because it means that, for a 45 Ohmoverall impedance requirement (as discussed hereinbefore with referenceto FIG. 1), the linear region of MOS transistor M1 (14) has to make upgreater than 50% of that total impedance (or 45 Ohms). Such a highimpedance requirement from transistor M1 (14) during its linear regionof operation may create an impedance curvature of as much as severalOhms over the 1.1V range of the USB2.0 specification (discussedhereinabove with reference to FIG. 1). If one chooses Rs 22 much higherthan 28 Ohms, then the dimensions (W/L) of transistor M1 (14) would beunrealistically huge as can be seen from equations (1) and (2). Anexternal precision resistor (not shown) of high value may be used (asdiscussed hereinbelow) to maintain the smaller size of M1, but therewould still be some linear region curvature (in the Ipad-Vpad graph) andalso a need for extra pads (for external precision resistor).

A problem with the current precision resistor approach is that manynon-analog CMOS fabrication processes do not have precision resistorsavailable. In such processes, it may not be possible to meet a tightimpedance specification with the circuit of FIG. 2. In such processes,some additional form of PVT impedance trimming is usually necessary. Atpresent, three basic solutions (or mixtures of them) exist to controlpad impedance within a tight percentage spread.

In the first approach, an external series precision resistor Rs (notshown) is used with essentially no PVT variation. The external resistoris not fabricated along with the MOS transistor M1 (14), but ratherexternally attached to the IC pad 24 in series with transistor M1 whenneeded. In this configuration, the internal resistor 22 may be absentfrom the MOS driver circuit 12. If such an external resistor Rs is madelarge enough, then one can effectively compensate for large PVTvariations in the impedance of transistor M1. Unfortunately, there aresome disadvantages with this technique: (1) This technique requiresexternal components and increases system cost and component count. (2)This technique may require more I/O pads (e.g., in case of using suchtechnique for a USB1.1 buffer in a USB2.0 PHY device). (3) Even with anexternal precision resistor, it may not be possible to meet a tight lowimpedance specification (e.g., under USB2.0 as discussed above withreference to FIG. 1) because of the need to make transistor M1inordinately large in size (as can be seen from equations (1) and (2)above).

In the second approach, a “digital” (or “discrete”) impedance trimmingtechnique is used. In this technique, transistor M1 (14) may comprise ofa number of NMOS “fingers,” the “correct” number of fingers beingactivated for any given PVT corner. The advantage of this technique isthat Vgs 17 can be kept constant, normally at its maximum possible value(e.g., supply voltage Vdd (not shown)), to minimize the curvature in thelinear region impedance of transistor M1 (14). However, thedisadvantages of this digital calibration technique are: (1) When it isnot possible to drive Vgs to a high value (>>Vds(max) in the Vpad regionof interest), then the linear region impedance curvature of transistorM1 (14) can still be very significant. When one takes into accountcomparator offsets and reference voltage offsets (in the trimmingcircuit realization), then a tight impedance specification may not berealizable. (2) In this technique, a whole replica pad driver isrequired for comparison with some reference precision resistor (orcurrent derived from such a precision resistor). This is a waste ofsilicon area, for example, in a USB2.0 PHY device that needs just twoimpedance controlled IC pads. The addition of an entire third pad forjust impedance control may not be desirable. (3) It is not clear underthis technique when one needs to update the impedance (i.e., change thenumber of NMOS “fingers”). Impedance updating may not be done safelyduring data transmission because it could cause major discontinuities inthe rise/fall in voltage/current characteristics of the transmittingpad. This may cause the overall system to fail. Therefore, the impedanceneeds to be updated at a time when there is no active data transmission(or data reception, if the pad is being used as a termination resistanceas, for example, in the USB2.0 HS mode). However, there may be no clear“gap” in the transmission protocol and, hence, choosing such “updategap” may pause an additional problem for the system designer. Such“update gaps” may be defined in the data protocol; however, that is notthe case for USB2.0. The USB2.0 system designer may have to spend timeand resource to figure out such “safe gaps” during which impedance maybe updated.

In the third approach, an “analog” impedance trimming technique is used.This technique essentially solves many of the problems associated withthe digital trimming technique described above. The advantage of theanalog trimming technique are: (1) A “fraction” or “scaled version” ofthe pad cell (i.e., a scaled replica pad driver) can be used forimpedance trimming. This saves on silicon area. (2) The trimming processis continuous; thus, one does not need a clear “update” gap in thetransmission protocol as with the digital solution. However, thedisadvantages of the classical analog solution are: (1) Similar todigital trimming, the analog approach trims the pad impedance for onevalue of Vpad only. This means that the linear region MOS impedancecurvature is not trimmed out for a range of Vpad values. (2) The use ofa properly-biased, scaled replica pad may require very tight control ofvoltage and current references because of the existence of MOS linearregion impedance curvature. (3) The Vgs of transistor M1 (14) cannot bekept constant, at its maximum possible value, to minimize linear regionimpedance curvature. With a fixed size for transistor M1, Vgs 17 will bemaximized for the slow process corner (slow process, maximumtemperature, minimum supply voltage Vdd, largest internal linearizingresistor Rs value) and, thus, the impedance curvature is minimized forthis worst case corner only. For the fast process corner (fast process,minimum temperature, maximum Vdd, smallest Rs value), however, the Vgs17 will have to be much lower than maximum, resulting in the worstlinear region impedance curvature (e.g., as shown in graph 30 in FIG.2). If the PVT change in Rs 22 is large, then the classic analogsolution fails to trim pad impedance to within a tight percentagetolerance.

Therefore, it is desirable to devise an analog solution (given its manyadvantages in terms of, for example, reduced complexity and conservationof silicon area) to trim out the MOS linear region impedance curvaturewhile accommodating PVT spreads in values of internal or externalprecision resistors. It is further desirable to maintain a linearimpedance for a range of Vpad values, while also maintaining the Vgs atits maximum possible value to obtain greater linearity.

SUMMARY

In one embodiment, the present disclosure contemplates a method ofcorrecting impedance curvature in a MOS driver circuit. The methodcomprises using a first MOS transistor and second MOS transistor as partof the MOS driver circuit; and operating the first MOS transistor andthe second MOS transistor so as to compensate for changes in outputimpedance of the first MOS transistor through corresponding changes inoutput impedance of the second MOS transistor.

In another embodiment, the present disclosure contemplates a method ofcorrecting impedance curvature in a MOS driver circuit. The methodcomprises using a first MOS transistor and second MOS transistor as partof the MOS driver circuit; and operating the first MOS transistor andthe second MOS transistor so as to increase output impedance of thesecond MOS transistor when output impedance of the first MOS transistordecreases, and vice versa.

In an alternative embodiment, the present disclosure contemplates amethod of correcting impedance curvature in a MOS driver circuit. Themethod comprises using a first MOS transistor and second MOS transistoras part of the MOS driver circuit; using a signal adder circuit as partof the MOS driver circuit; maintaining a controlled voltage at a firstinput terminal of the first MOS transistor; using the signal addercircuit to provide a differential voltage at a second input terminal ofthe second MOS transistor; and operating the first MOS transistor andthe second MOS transistor so as to compensate for changes in outputimpedance of the first MOS transistor through corresponding changes inoutput impedance of the second MOS transistor.

In a further embodiment, the present disclosure contemplates a method ofcorrecting impedance curvature in a MOS driver circuit. The methodcomprises using a first MOS transistor and second MOS transistor as partof the MOS driver circuit; using a signal adder circuit as part of theMOS driver circuit; maintaining a controlled voltage at a first inputterminal of the first MOS transistor; using the signal adder circuit toprovide a differential voltage at a second input terminal of the secondMOS transistor; and operating the first MOS transistor and the secondMOS transistor so as to increase output impedance of the second MOStransistor when output impedance of the first MOS transistor decreases,and vice versa.

In another embodiment, the present disclosure contemplates a MOS drivercircuit that comprises a first MOS transistor configured to receive acontrolled voltage at a first input terminal thereof; and a second MOStransistor coupled to the first MOS transistor and configured to receivea differential voltage at a second input terminal thereof, wherein thesecond MOS transistor is configured to have an increased outputimpedance when output impedance of the first MOS transistor decreases,and vice versa.

In a still further embodiment, the present disclosure contemplates a MOSdriver circuit that comprises a first MOS transistor configured toreceive a controlled voltage at a first input terminal thereof; and asecond MOS transistor coupled to the first MOS transistor and configuredto receive a differential voltage at a second input terminal thereof,wherein the second MOS transistor is configured to compensate forchanges in output impedance of the first MOS transistor throughcorresponding changes in output impedance of the second MOS transistor.

The present disclosure further contemplates a MOS driver circuit thatcomprises a first MOS transistor configured to receive a controlledvoltage at a first input terminal thereof; and a second MOS transistorcoupled to the first MOS transistor and configured to receive adifferential voltage at a second input terminal thereof, wherein thesecond MOS transistor is configured to have an increased outputimpedance when output impedance of the first MOS transistor decreases,and vice versa; a signal adder circuit coupled to the first and thesecond MOS transistors, wherein an output of the signal adder circuit iscoupled to the second input terminal to provide the differential voltageto the second MOS transistor, and wherein a first output terminal of thefirst MOS transistor is coupled to a first input of the signal addercircuit to provide a first input voltage thereto; a scaled replica ofthe MOS driver circuit having an output coupled to a second input of thesignal adder circuit to provide a second input voltage thereto; and alinearizing resistor coupled in series with the first output terminal ofthe first MOS transistor and a second output terminal of the second MOStransistor.

The present disclosure still further contemplates a MOS driver circuitthat comprises a first MOS transistor configured to receive a controlledvoltage at a first input terminal thereof; a second MOS transistorcoupled to the first MOS transistor and configured to receive adifferential voltage at a second input terminal thereof, wherein thesecond MOS transistor is configured to compensate for changes in outputimpedance of the first MOS transistor through corresponding changes inoutput impedance of the second MOS transistor; a signal adder circuitcoupled to the first and the second MOS transistors, wherein an outputof the signal adder circuit is coupled to the second input terminal toprovide the differential voltage to the second MOS transistor, andwherein a first output terminal of the first MOS transistor is coupledto a first input of the signal adder circuit to provide a first inputvoltage thereto; a scaled replica of the MOS driver circuit having anoutput coupled to a second input of the signal adder circuit to providea second input voltage thereto; and a linearizing resistor coupled inseries with the first output terminal of the first MOS transistor and asecond output terminal of the second MOS transistor.

In a further embodiment, the present disclosure contemplates a method ofoperating a MOS driver circuit. The method comprises using a first MOStransistor with a first terminal, a second terminal, and a thirdterminal; using a second MOS transistor with a fourth terminal, a fifthterminal, and a sixth terminal; providing a DC supply voltage to thefirst and the fourth terminals; further providing a reference potentialto the second and the fifth terminals; and further using an internallinearizing resistor and an external precision resistor in series withthe third and the sixth terminals.

In a still further embodiment, the present disclosure contemplates asystem that comprises a processor; a memory controller; a memory device;a first bus interconnecting the processor and the memory controller; anda second bus interconnecting the memory controller and the memorydevice, wherein at least one of the processor, the memory controller,and the memory device includes a MOS driver circuit. The MOS drivercircuit comprises a first MOS transistor configured to receive acontrolled voltage at a first input terminal thereof, and a second MOStransistor coupled to the first MOS transistor and configured to receivea differential voltage at a second input terminal thereof, wherein thesecond MOS transistor is configured to compensate for changes in outputimpedance of the first MOS transistor through corresponding changes inoutput impedance of the second MOS transistor.

The present disclosure also contemplates a system that comprises a dataprocessing unit; an input device connected to the data processing unit;an output device connected to the data processing unit; and a datastorage device connected to the data processing unit. The dataprocessing unit includes a processor, a memory controller, a memorydevice, a first bus interconnecting the processor and the memorycontroller, and a second bus interconnecting the memory controller andthe memory device. At least one of the processor, the memory controller,the memory device, the input device, the output device, and the datastorage device includes a MOS driver circuit, which comprises a firstMOS transistor configured to receive a controlled voltage at a firstinput terminal thereof, and a second MOS transistor coupled to the firstMOS transistor and configured to receive a differential voltage at asecond input terminal thereof, wherein the second MOS transistor isconfigured to have an increased output impedance when output impedanceof the first MOS transistor decreases, and vice versa.

The present disclosure describes a system and method to correct orcancel MOS linear region impedance curvature. The linear regioncurvature correction may be obtained by using two MOS transistors in thepad driver/buffer and operating the transistors so as to proportionatelyincrease output impedance of one of the transistors when the outputimpedance of the other decreases, and vice versa. A linear pad impedancemay be maintained over a range of Vpad values, while also maintainingthe Vgs supplied to pad driver transistors at its maximum possible valueto obtain greater linearity. The approach of the present disclosurerelaxes the requirements on the voltage/current references used in theMOS pad drivers and makes tight impedance control possible, especiallyin a situation where the MOS fabrication process (typically allcurrently used processes) does not have available an internal precisionresistor with reasonably well controlled value. Thus, PVT spreads invalues of internal or external precision resistors are also accommodatedin a MOS driver circuit built according to the teachings of the presentdisclosure.

BRIEF DESCRIPTION OF THE DRAWINGS

For the present disclosure to be easily understood and readilypracticed, the present disclosure will now be described for purposes ofillustration and not limitation, in connection with the followingfigures, wherein:

FIG. 1 illustrates an exemplary pad impedance specification under USB2.0for a low impedance driver circuit;

FIG. 2 illustrates a general approach to pad impedance control in a MOSdriver circuit;

FIG. 3 illustrates an exemplary MOS driver circuit for trimming MOSlinear region impedance curvature according to one embodiment of thepresent disclosure;

FIG. 4 is an exemplary schematic of a practical realization of the MOSdriver circuit in FIG. 3 according to one embodiment of the presentdisclosure;

FIGS. 5A and 5B illustrate two diagrams showing plots of voltages andcurrents (versus time), respectively, at various terminals or points inthe MOS driver circuit in FIG. 4 when the circuit is SPICE simulated inthe slow SS corner, with rising edge for Vpad;

FIGS. 6A and 6B illustrate two diagrams showing plots of voltages andcurrents (versus time), respectively, at various terminals or points inthe MOS driver circuit in FIG. 4 when the circuit is SPICE simulated inthe fast (FF) corner, with rising edge for Vpad;

FIGS. 7A and 7B illustrate two diagrams showing zoomed (closer) versionsof respective voltage and current plots in FIGS. 6A-6B;

FIGS. 8A and 8B illustrate two diagrams showing plots of voltages andcurrents (versus time), respectively, at various terminals or points inthe MOS driver circuit in FIG. 4 when the circuit is SPICE simulated inthe fast (FF) corner, with falling edge for Vpad;

FIG. 9 is an exemplary schematic of a practical realization of the MOSdriver circuit in FIG. 3 according to one embodiment of the presentdisclosure;

FIGS. 10A and 10B illustrate two diagrams showing plots of voltages andcurrents (versus time), respectively, at various terminals or points inthe MOS driver circuit in FIG. 9 when the circuit is SPICE simulated inthe slow SS corner, with rising edge for Vpad;

FIGS. 11A and 11B illustrate two diagrams showing plots of voltages andcurrents (versus time), respectively, at various terminals or points inthe MOS driver circuit in FIG. 9 when the circuit is SPICE simulated inthe fast (FF) corner, with rising edge for Vpad;

FIG. 12 is an exemplary schematic of a practical realization of the MOSdriver circuit in FIG. 3 according to one embodiment of the presentdisclosure;

FIGS. 13A-13C illustrate three diagrams showing plots of voltages andcurrents (versus time) at various terminals or points in the MOS drivercircuit in FIG. 12 when the circuit is SPICE simulated in the slow SScorner, with rising edge for Vpad;

FIGS. 14A-14C illustrate three diagrams showing plots of voltages andcurrents (versus time) at various terminals or points in the MOS drivercircuit in FIG. 12 when the circuit is SPICE simulated in the fast (FF)corner, with rising edge for Vpad;

FIGS. 15A and 15B illustrate slow (SS) corner results of SPICEsimulation of the MOS driver circuit in FIG. 4 in the HS mode;

FIGS. 16A and 16B illustrate fast (FF) corner results of SPICEsimulation of the MOS driver in FIG. 4 in the HS mode;

FIG. 17 illustrates how, when a bandgap circuit operates on a supplyother than the 3.3V used for a pad driver circuit according to thepresent disclosure (e.g., on a 1.5V supply), one can effectively “levelshift” the bandgap voltage to generate the required high and low valuesfor the reference voltage (Vref) to a MOS pad driver circuit constructedaccording to teachings of the present disclosure;

FIG. 18 illustrates the MOS driver circuit in FIG. 3 with the currentreference “Iref” supplied by a separate bias generator so as to allowthe use of the driver circuit as a stand alone USB1.1 buffer;

FIG. 19 illustrates a MOS driver circuit, which is identical to MOSdriver in FIG. 3, but with its impedance control inhibited so as toallow the use of the driver circuit as a stand alone USB1.1 buffer; and

FIG. 20 is a block diagram depicting a system in which a MOS drivercircuit constructed according to the teachings of the present disclosuremay be used.

DETAILED DESCRIPTION

Reference will now be made in detail to certain embodiments of thepresent disclosure, examples of which are illustrated in theaccompanying drawings. It is to be understood that the figures anddescriptions of the present disclosure included herein illustrate anddescribe elements that are of particular relevance to the presentdisclosure, while eliminating, for the sake of clarity, other elementsfound in typical pad drivers or buffers. It is noted at the outset thatthe terms “connected”, “coupled,” “connecting,” “electricallyconnected,” etc., are used interchangeably herein to generally refer tothe condition of being electrically connected. Further, a terminal isconsidered “held” at a specific potential when appropriate voltage(e.g., ground potential, reference voltage, supply voltage, etc.) isavailable at that terminal. The terms “supply” or “provide” are alsoused interchangeably to refer to electrically supplying or providing avoltage or current to a circuit element or terminal.

It is further noted at the outset that the term “MOS”, as used herein,includes a wide range of semiconductor devices. Some of the MOS devices(e.g., transistors) may have at least three terminals, whereas someother MOS devices (e.g., resistors or diodes) may have less than threeterminals. In one embodiment, these devices are fabricated using a CMOS(complementary MOS) fabrication technology. A MOS device can include,for example, a Field Effect Transistor (FET), a dynamic-threshold MOStransistor (DTMOST), or a BICMOS (bipolar CMOS) device. The FET can be aMetal Oxide Semiconductor FET (MOSFET), a Junction FET (JFET), or aMetal Semiconductor FET (MESFET). A “three terminal device” can includedevices with three or more terminals. For example, a three terminaldevice can be an FET with a fourth terminal (a bulk, backgate orsubstrate electrode) coupled to a voltage potential. In one embodiment,the bulk electrode, when used, is coupled to a ground potential or tothe source electrode for an n-channel MOSFET (or NMOS), or to the sourceelectrode or the positive supply rail for a p-channel MOSFET (or PMOS)as will be evident from the circuit diagrams.

FIG. 3 illustrates an exemplary MOS driver circuit 32 for trimming MOSlinear region impedance curvature according to one embodiment of thepresent disclosure. The MOS driver circuit 32 uses analog impedancetrimming techniques and may be used in place of the conventional MOSdriver circuit 12 (FIG. 2) to obtain linear pad resistance (Rpad) asdiscussed hereinbelow. It is noted here that the terms “impedance” and“resistance” are generally used interchangeably herein, unless otherwisespecified. The MOS driver circuit 32 may comprise a pad driver 34 and apad driver replica 36, which may be a scaled replica (as is illustrated,for example, in FIG. 4 discussed later). Various circuit connections inthe driver circuit 32 are self-explanatory and, hence, are very brieflydescribed herein.

The pad driver 34 may include two NMOS transistors Ma 38 and Mb 42 withtheir respective source terminals 40, 44 held at a common circuitpotential (e.g., the circuit ground), and their respective drainterminals 41, 45 connected to each other and also in series with theinternal linearizing resistor Rs 22. A controlled voltage Vgs ismaintained at the gate terminal 39 of the transistor Ma 38 (via anamplifier 60 discussed later below), whereas a differential voltage(Vds−Vds_ref) is provided to the gate terminal 43 of the transistor Mb42 via a voltage adder or summer circuit 46 discussed below. In oneembodiment, the signal adder circuit 46 is an operational (differential)amplifier with non-inverting input 48 connected to the drain terminal 41of transistor Ma 38 and inverting input 47 connected to the drainterminal of transistor M1 (52) in the pad driver replica 36 as shown.The output 49 of the voltage summer 46 may be connected to the gate 43of transistor Mb 43 to supply the differential voltage at the gate 43. Asuitable feedback load 50 may be placed between the output 49 andinverting input 47 of the op-amp 46 as shown. The pad driver circuit 34is connected to the IC pad 24, which allows coupling with other circuitelements (not shown) during operation of a system. The presence of thepad voltage Vpad 26 is also indicated in FIG. 3.

The pad driver replica 36 may include an NMOS transistor M1 (52) havinga source terminal 54 connected to the circuit ground, a gate terminal 53connected to an output 63 of an inverting amplifier 60, and a drainterminal 55 connected to a current reference source 58 and a positivesupply voltage Vdd 59 via a series resistor 56. In the embodiment ofFIG. 3, the value of the series resistor 56 is chosen to be 2m times thevalues of the internal linearizing resistor Rs 22, where “m” is a numberequivalent to the number of transistors M1 (52) (i.e., the NMOS“fingers”) connected in parallel to make up the pad driver transistorsMa 38 and Mb 42 as is indicated in FIG. 3. For example, in theembodiment of FIG. 4, the value of m=18 (i.e., the number of“fingers”=18) as can be seen from the pad driver circuit portion in FIG.4 (only the value of “m” is given in FIG. 4, without depicting all 18fingers).

The MOS driver circuit 32 may also include an amplifier 60 configured tomaintain the controlled voltage Vgs at the gates of transistors M1 (52)and Ma 38 as is shown in FIG. 3. The amplifier 60 may be a differentialamplifier operating in an inverting configuration with its output 63connected to the gates 53 and 39, its inverting input 61 connected tothe output of the current reference 58 (and, hence, also indirectly tothe drain 55 of transistor M1) to receive appropriate portion of thesupply voltage Vdd, and its non-inverting input 62 held at apredetermined reference potential Vref. The differential invertingamplifier 60 thus maintains a constant, steady potential Vgs at thegates of replica driver transistor M1 (52) and the pad driver transistorMa 38. This allows the value of Vgs to be continually updated tomaintain the impedance of M1 (52) and Ma 38 constant with PVT (process,voltage, temperature) variations. Thus, the amplifier 60 ensures thatcontinuous analog trimming of Vgs is performed. The scaled impedance ofthe pad driver replica 36 may be exactly equal to that of the pad driver34, only when the voltage of pad 24 is at Vref. However, in the event ofpad voltage above and below Vref, the teachings of the presentdisclosure may be used to maintain impedance of the pad driver 34 nearto that of the replica 36 (which is held constantly at Vref).

In the embodiment of FIG. 3, the voltage appearing at the invertinginput 61 of the amplifier 63 is denoted as Vref_fb (the feedback portionof the reference voltage Vref), the voltage between the drain terminal55 of transistor M1 (52) and the ground potential 57 is denoted asVds_ref (which is also an input to the voltage summer 46), the identicalvoltages at the gate terminals 53, 39 of transistors M1 (52) and Ma 38respectively are commonly denoted as Vgs, and the voltage between thedrain and source terminals of transistor Mb 42 is denoted as Vds. It isobserved that as the drains of transistors Ma and Mb are connected, thesame voltage Vds is also present between the drain 41 and source 40 oftransistor Ma. Further, as the drain of transistor Ma 38 is connected tothe non-inverting input 48 of the op-amp 46, the op-amp 46 receivesvoltage Vds at its non-inverting input 48. Hence, the differentialvoltage at the output 49 of the op-amps 46 may be given as Vds−Vds_ref.If the gate-to-source voltage of transistor Mb 42 is denoted as Vgsb,then the effective voltage at the gate terminal 43 of transistor Mb 42may be given as Vgsb=Vgs−Vds_ref+Vds, which includes a differentialvoltage component (Vds−Vds_ref) as can be seen from the discussionherein. The voltage Vgsb will be equal to Vgs only when the voltage atthe IC pad 24 is exactly equal to Vds_ref, because, in that event, thenon-inverting input of the op-amp 46 will be at Vds_ref.

Assuming proper biasing of all circuit elements in FIG. 3, in operation,the amplifier 60 sets the correct (desired) Vgs for M1 (52) (and, hence,for Ma 38) such that Vref_fb is at the same potential as Vref. In oneembodiment, Vref=1.1V. When the voltage on the IC pad 24 (i.e., Vpad 26)is driven to Vref (i.e., Vpad=Vref), the value of Ipad equals to2*m*Iref. Hence, the differential voltage output 49 of the adder circuit46 is zero and, therefore, the gate voltages of transistors Ma 38 and Mb42 become identical (=Vgs). On the other hand, when the voltage on theIC pad 24 is driven below Vref (i.e., Vpad<Vref), the resistance RMa(not shown) (i.e., impedance of transistor Ma 38) reduces with reducingVds, which reduces with reducing Vpad (because Vds=Vpad−Ipad*Rs).However, the resistance RMb (not shown) (i.e., impedance of transistorMb 42) increases with reducing Vds by an amount exactly equal to thereduction in RMa. Therefore, the net effect is a linear pad resistanceas can be seen from the combined linear region Ipad-Vpad graph 66 forthe pad driver 32 with MOS transistors Ma 38 and Mb 42. The linearity isthus maintained for Vpad values less than or equal to the referencevoltage Vref. The following equations further explain why RMb increasesby the same rate as RMa is decreasing.

It is seen from equation (1) that, in general, the resistance for MOS inthe linear region can be given by:

$\begin{matrix}{R_{MOS} = \frac{1}{\beta\left( {V_{gs} - V_{T} - \frac{V_{ds}}{2}} \right)}} & (3)\end{matrix}$All the parameters in equation (3) are explained before with referenceto equations (1) and (2). Based on the general equation (3) and variousvoltages shown in FIG. 3, the values of RMa and RMb can be given as:

$\begin{matrix}{R_{Ma} = \frac{1}{\beta\left( {V_{gs} - V_{T} - \frac{V_{ds}}{2}} \right)}} & (4) \\{R_{Mb} = {\frac{1}{\beta\begin{pmatrix}{\left\lbrack {V_{gs} - V_{ds\_ ref} + V_{ds}} \right\rbrack -} \\{V_{T} - \frac{V_{ds}}{2}}\end{pmatrix}} = \frac{1}{\beta\begin{pmatrix}{V_{gs} - V_{ds\_ ref} -} \\{V_{T} + \frac{V_{ds}}{2}}\end{pmatrix}}}} & (5)\end{matrix}$It can be seen from equations (4) and (5) that the value of RMadecreases as Vds goes lower than Vds_ref, but the value of RMb increasesby the same rate (as RMa is decreasing) when Vds goes below Vds_ref.Therefore, the combined effective resistance of RMa and RMb is linearover a range of Vpad values as can be seen from the Ipad-Vpad graph 66in FIG. 3.

On the other hand, the vice versa is also true. That is, when thevoltage on the IC pad 24 is driven above Vref (i.e., Vpad>Vref), theresistance RMa (not shown) increases with increasing Vds, but theresistance RMb (not shown) decreases with increasing Vds by an amountexactly equal to the reduction in RMa. Therefore, the net effect is alinear pad resistance similar to that illustrated in the graph 66, butwith curves of RMa and RMb reversed (not shown in FIG. 3). Thus, thelinearity may also be maintained for Vpad values greater than thereference voltage Vref. It is noted, however, that Vpad may not begreater than Vref in USB2.0 circuits.

A “pad” or “integrated circuit pad” (defined hereinbefore) may be usedfor inter-system communication, for example, via a cable of controlledcharacteristic impedance. It is important, in such an environment, totightly control transmitter and receiver pad impedances, in order toavoid multiple signal reflections. Therefore, the transmitting IC pad(having the impedance control according to the methodology of thepresent disclosure) may connect to a terminating receiver pad, which maybe within the same IC or in another external IC. This connection may bedirect, via a cable (possibly with intermediate passive components, likeseries resistors or inductors), or via any other suitable manner toavoid multiple signal reflections.

Many possible architectures may exist for the voltage summer circuit 46if only the DC resistance (Rpad_dc) of the IC pad 24 is of concern. Incase of a USB1.1 buffer with the MOS driver circuit 32, for example, thepad impedance (Rpad) should be within USB1.1 specification when the padis subjected to a return wave-front from an unterminated receiver (notshown) connected to the IC pad 24. In one embodiment, the transmittedwave slew rate may be set so as to achieve a rise/fall time of 20 ns orless for the return wave-front. To achieve such high-speed switchingwith desired Rpad, the summer circuit 46 may be designed as a voltagefollower circuit as is illustrated, for example, in FIGS. 4, 9 and 12.

First Circuit Realization of FIG. 3

FIG. 4 is an exemplary schematic of a practical realization of the MOSdriver circuit 32 in FIG. 3 according to one embodiment of the presentdisclosure. The “compensated” pad driver circuit (or MOS driver circuit)68 in FIG. 4 is shown along with a conventional “uncompensated” paddriver 72 (similar to the driver circuit 12 in FIG. 2) just forcomparison purpose. As FIG. 4 is a realization of the general circuitconfiguration in FIG. 3, the circuit blocks having similar functions inFIGS. 3 and 4 are denoted by the same reference numerals withoutassigning reference numerals to each circuit component in FIG. 4 (e.g.,resistor, transistor, capacitor, etc.). Further, as the function of thecircuit in FIG. 4 can be easily explained when a comparison of FIGS. 3and 4 is made, a detailed description of the circuit components in FIG.4 is omitted here for the sake of brevity. From a comparison of FIGS. 3and 4, it is seen that the MOS driver circuit 68 in FIG. 4 includes thepad driver 34, the scaled replica of the pad driver 36, the voltagesummer circuit 46 (implemented in a voltage follower configuration) andan amplifier 70. The amplifier 70 is substantially similar to theamplifier 60 in FIG. 3, except that the amplifier 70 is configured toprovide a bias for the follower circuit 46 instead of a direct drainconnection (connecting drain 55 with the inverting input 47) in FIG. 3.The follower bias line is indicated by the reference numeral 71. It isobserved that the Vref_fb signal is provided to the amplifier 70 whenthe test pads 73 and 74 are connected. Despite some implementationalvariations, the driver circuit 68 in FIG. 4 is functionally equivalentto the circuit 32 in FIG. 3.

Some points to note about the MOS driver circuit 68 of FIG. 4 are: (1) Avoltage follower circuit is included at the output of the amplifier 70so as to automatically generate the correct bias for the voltage summercircuit 46 (also implemented as a voltage follower) in the pad driver34. Thus, the amplifier 70 includes a differential amplifier and avoltage follower. (2) Ideally, one would want the heavy capacitive loadon node ‘y’ (designated by reference numeral “76” in FIG. 4) tocompensate the amplifier 70. In order to make node ‘y’ the dominantpole, one could use MOS diode connections for both PMOS loads(transistors M61 and M62 in FIG. 4) of the differential amplifier in theamplifier 70. However, then there may not be enough overall voltage gainfor the amplifier 70 and the input offset voltage (for the amplifier 70)may be too large (creating unacceptable errors in the output impedanceof the MOS driver circuit 68). (3) The 5 pF compensation capacitor C0(78) may be dramatically reduced in value (or even eliminated) ifself-biased cascode stages were used for the PMOS loads of thedifferential amplifier. (4) The 0.1 Ohm resistors 79-81 are used tofacilitate current measurement for SPICE simulations. They are not‘real’ resistors in the sense that they may not be part of the finalcircuit as fabricated. (5) As noted before, the uncompensated padconnection 72 is included in FIG. 4 for simulation comparison purposesonly. (6) There may be a relatively large change in the follower biascurrent (i.e., large changes in the current in MOS transistors M65 andM66) between the slow corner case (largest bias current) and the bestcorner case (smallest bias current). This may be desirable because theincreased bias current in the slow corner may compensate for increasedNMOS gate capacitance of transistor M103. The percentage change may be,however, much more than necessary to compensate for increasedcapacitance. (7) Because the output of the differential amplifier (inamplifier 70) may not drive down near to 0V, the top PMOS (transistorM63) of the follower circuit (in amplifier 70) may have to be largerthan it would otherwise need to be. In this event, a two stage amplifier(with an inherent level shift, capable of driving near to 0V) may be abetter solution as depicted, for example, in the embodiment of FIG. 12.(8) The number of NMOS fingers (m=18, transistors M102 and M103) in thepad driver 34 are chosen to maximize the gate drive voltage (Vgs) in theslow corner (slow process, maximum temperature, minimum Vdd, largest Rsvalue). When the pad driver circuit 34 is then subjected to the fastcorner (fast process, minimum temperature, maximum Vdd, smallest Rsvalue), the NMOS Vgs gate drive voltage will be as high as possible.Hence, such a scheme of maximizing Vgs voltage keeps the MOS linearregion curvature to a minimum; however, it will not completely eliminatethe curvature.

SPICE Simulation Results for the Circuit 68 of FIG. 4

FIGS. 5A and 5B illustrate two diagrams 94, 98 showing plots of voltagesand currents (versus time), respectively, at various terminals or pointsin the MOS driver circuit 68 in FIG. 4 when the circuit 68 is SPICEsimulated in the slow SS corner, with rising edge for Vpad. The voltagesin diagram 94 are measured in volts and mV, whereas the currents indiagram 98 are measured in mA. The time is given in nanoseconds in bothdiagrams. Various plots or graphs in the voltage diagram 94 arereferenced by the same numerals as those used to identify correspondingtest locations in the circuit in FIG. 4 where the voltage measurementsare taken. For example, the voltage plot 86 corresponds to the voltagemeasurements taken at location 86 (the “x” signal line) in FIG. 4.Similarly, the voltage plot 87 corresponds to the voltage measurementstaken at location 87 (the “fb_int_comp” line) in FIG. 4, and so on. Inthe same manner, the current diagram 98 includes a number of plots(except plots 91, 93, 95) bearing reference numerals corresponding tothe locations of current measurements in FIG. 4. For example, thecurrent graph 90 corresponds to the current measurements taken at the“padd” terminal 90 in FIG. 4 (similar to the IC pad 24 in FIG. 3),thereby taking the Rpad into account. Similarly, the current plot 92corresponds to the current measurements taken at the “padd-unc” terminal92 in FIG. 4, thereby taking the Rpad of the uncompensated driver 72into account. It is noted that the plot 91 defines a current plot for anideal resistance of 40.5 Ohms, the plot 93 defines a current plot for anideal resistance of 49.5 Ohms, and the plot 95 defines an ideal currentof 22 mA. These resistor values, 40.5 Ohms and 49.5 Ohms, respectivelycorrespond to the lower and upper limits on the pad impedance valuesgiven in FIG. 1 for the USB2.0 impedance specification. The idealcurrent of 22 mA corresponds to the upper 49.5 Ohms limit when the padvoltage is 1.09V (˜1.1V) as can be seen from FIG. 1. It is observed herethat these ideal values are not part of the MOS driver circuit 68 inFIG. 4. These values are plotted in FIG. 5B so as to visualize whether apad design is meeting the USB2.0 impedance specification.

It is seen from the voltage and current plots in FIGS. 5A-5B for theslow SS corner (worst for MOS drive capability) that the pad driver NMOSVgs is maximized (>2.8V from a 3V supply) as can be seen from the plot76. It is also seen from a comparison of current plots 90, 92 that thereis no real discernible difference between the performance of curvaturecompensated and uncompensated pads in FIG. 4. Therefore, curvaturecompensation may not be needed in the slow (SS) corner case.

FIGS. 6A and 6B illustrate two diagrams 102, 104 showing plots ofvoltages and currents (versus time), respectively, at various terminalsor points in the MOS driver circuit 68 in FIG. 4 when the circuit 68 isSPICE simulated in the fast (FF) corner, with rising edge for Vpad. Thevoltages in diagram 102 are measured in volts and mV, whereas thecurrents in diagram 104 are measured in mA. The time is given innanoseconds in both diagrams. Various plots or graphs in the diagrams102, 104 are referenced in the same manner (including the plots 91, 93,and 95 for ideal resistor and current values) as that described abovewith reference to FIGS. 5A-5B. Therefore, no additional explanation ofvarious plots in FIGS. 6A-6B is provided. As can be seen from plot 76 inFIG. 6A, in the fast FF corner (best for MOS drive capability), paddriver NMOS Vgs has dropped to just 1.3V. This leads to several Ohms oflinear region MOS impedance curvature. When one takes into accountpercentage errors in the generation of Vref and Iref, it may bedifficult to meet the 45 Ohm ±10% impedance specification (under USB2.0described before) using the uncompensated pad 72. It can be seen from acomparison of plots 90 and 92 that linear region curvature compensationis desirable in the fast (FF) corner case.

FIGS. 7A and 7B illustrate two diagrams 106, 108 showing zoomed (closer)versions of respective voltage and current plots 102, 104 in FIGS.6A-6B. As before, the plots 106, 108 of voltages and currents (versustime), respectively, are obtained at various terminals or points in theMOS driver circuit 68 in FIG. 4 when the circuit 68 is SPICE simulatedin the fast (FF) corner, with rising edge for Vpad. As FIGS. 7A-7B arezoomed versions of corresponding FIGS. 6A-6B, no additional explanationof graphs in FIGS. 7A-7B is provided. It is, however, observed from thediagram 108 that the compensated circuit 68 according to the presentdisclosure provides a very linear response (plot 90) as compared to therelatively large curvature (plot 92) for uncompensated driver circuit72.

FIGS. 8A and 8B illustrate two diagrams 110, 112 showing plots ofvoltages and currents (versus time), respectively, at various terminalsor points in the MOS driver circuit 68 in FIG. 4 when the circuit 68 isSPICE simulated in the fast (FF) corner, with falling edge for Vpad. Thevoltages in diagram 110 are measured in volts and mV, whereas thecurrents in diagram 112 are measured in mA. The time is given innanoseconds in both diagrams. Various plots or graphs in the diagrams110, 112 are referenced in the same manner as that described above withreference to FIGS. 5A-5B. Therefore, no additional discussion of variousplots in FIGS. 8A-8B is provided.

It is noted that in various current graphs in FIGS. 5B, 6B, 7B, and 8B,the follower bias currents (in the MOS transistors M65 and M66 in FIG.4) are not shown. These bias currents range from 120 μA for the bestcase corner to 1.3 mA for the worst case corner. In the embodiment ofFIG. 4, because the switching is in the order of 10 mA plus during padslew (for Vpad), such bias currents would be acceptable. In the slewrate limiting circuits (not shown in FIG. 4) of a USB1.1 buffer, afollower similar to the follower 46 in FIG. 4, with its high biascurrent, may also be needed. It is observed here that although thecurrent source Iref 58 (FIG. 3) may preferably be an ideal currentsource, it may be replaced with a non-ideal, “real” current sourcewithout making any substantial difference to stability performance. TheVds of the real current source may be large (almost 2V in the worstcase) and, therefore, the PMOS (not shown) forming the non-ideal currentsource can be relatively small and therefore low capacitive.

Second Circuit Realization of FIG. 3

FIG. 9 is an exemplary schematic of a practical realization of the MOSdriver circuit 32 in FIG. 3 according to one embodiment of the presentdisclosure. The “compensated” pad driver circuit (or MOS driver circuit)114 in FIG. 9 is shown along with a conventional “uncompensated” paddriver 72 (similar to the driver circuit 12 in FIG. 2) just forcomparison purpose. As FIG. 9 is a realization of the general circuitconfiguration in FIG. 3, the circuit blocks having similar functions inFIGS. 3 and 9 are denoted by the same reference numerals withoutassigning reference numerals to each circuit component in FIG. 9 (e.g.,resistor, transistor, capacitor, etc.). Further, as the function of thecircuit in FIG. 9 can be easily explained when a comparison of FIGS. 3and 9 is made, a detailed description of the circuit components in FIG.9 is omitted here for the sake of brevity. From a comparison of FIGS. 3and 9, it is seen that the MOS driver circuit 114 in FIG. 9 includes thepad driver 34, the replica of the pad driver 36, the voltage summercircuit 46 (implemented in a voltage follower configuration), theinverting amplifier 60, and an additional amplifier 118. The amplifier118 is an additional amplifier connected between the replica pad driver36 and the voltage summer 46 to provide a bias for the follower circuit46 on the line indicated by reference numeral “119” in FIG. 9. Thus, theinverting input 47 (FIG. 3) to the voltage summer 46 is not provideddirectly from the pad driver replica 36, but through the amplifier 118.It is observed that the Vref_fb signal is provided to the amplifier 60when the test pads 120 and 121 are connected. Despite someimplementational variations, the driver circuit 114 in FIG. 9 isfunctionally equivalent to the circuit 32 in FIG. 3.

The MOS driver circuit 114 of FIG. 9 may be used, with its PMOS inputstage (in amplifier 60), when a low reference voltage (e.g., Vref=0.5V)is available. However, in the simulation of circuit 114, the value of1.1V was used for Vref. Some points to note about the MOS driver circuit114 of FIG. 9 are: (1) As noted before, a separate (2nd) amplifier 118is used to generate the bias for the follower circuit 46. The amplifier118 is shown ‘unbiased’, but a biased version could be used, if the NMOS(any of the NMOS in the circuit 114) V_(T) is not too great. All NMOS inthe circuit 114 may have V_(T) values that change with process. Theprimary amplifier 60 and a follower bias amplifier 118 combination maybe used so long as the improvement in MOS linear region linearity is notoutweighed by the double offset of these two amplifiers 60, 118. (2)Ideally, the heavily capacitively loaded NMOS gate of the pad drivertransistor M102 (FIG. 9) may compensate the amplifier 60 (dominantpole). However, in practice, the gate-drain capacitance of the paddriver NMOS (transistor M102) may create a significant amount of chargeinjection onto the gate node (of transistor M102) during pad switching.This may cause the gate node voltage to move considerably, necessitatingthe addition of a ‘slugging’ capacitor (capacitor C4 in FIG. 9).Generally, the voltage Vgs of transistor M102 should not change as thepad voltage switches; it (Vgs of M102) should preferably only changevery slowly with PVT variations. However, the Vgs of M102 may move fromits ideally static bias condition. For example, if the pad voltage isrising, then the capacitance between the drain and gate of M102 willtend to pull the Vgs node up, potentially by several 10's of mV. Theimpedance of M102 will therefore change. On the other hand, if the padvoltage is falling, then the capacitance between the drain and gate ofM102 will tend to pull the Vgs node down, potentially by several 10's ofmV. The impedance of M102 will again change. The amount of change in Vgsdepends on the capacitive divider given by the following approximation:ΔVgs=ΔVpad*(C _(drain-gate of M102))/(C _(on node 123 in FIG. 9))  (6)

Therefore, the capacitor C4 of 10 pF is added on node 123 (FIG. 9) tominimize ΔVgs. There may however be more elaborate, less space-costly(i.e., silicon area) solutions, to this problem including, for example,use of a 2 stage push-pull amplifier (not shown) with a high UGBW (UnityGain BandWidth) in place of the amplifier 60. (3) With this particularsingle stage differential amplifier circuit 60, it may be difficult todrive the gate of the NMOS pad driver (transistor M102) much above about2V. Therefore, it may be necessary to oversize the driver NMOS(transistor M102) to cope with the worst case SS corner. This problem,however, may be overcome for the particular realization in FIG. 9, byreplacing the PMOS input differential amplifier 60 with an NMOS inputdifferential amplifier such as, for example, the amplifier 70 used inthe circuit of FIG. 4. However, such an NMOS differential amplifier mayrequire the reference voltage (Vref) at least a volt higher than thatrequired by the PMOS amplifier.

SPICE Simulation Results for the Circuit 114 of FIG. 9

FIGS. 10A and 10B illustrate two diagrams 130, 134 showing plots ofvoltages and currents (versus time), respectively, at various terminalsor points in the MOS driver circuit 114 in FIG. 9 when the circuit 114is SPICE simulated in the slow SS corner, with rising edge for Vpad. Thevoltages in diagram 130 are measured in volts and mV, whereas thecurrents in diagram 134 are measured in mA. The time is given innanoseconds in both diagrams. Various plots or graphs in the voltagediagram 130 are referenced by the same numerals as those used toidentify corresponding test locations in the circuit in FIG. 9 where thevoltage measurements are taken. For example, the voltage plot 126corresponds to the voltage measurements taken at location 126 (the“ds_comp” signal line in the pad driver 34) in FIG. 9. Similarly, thevoltage plot 124 corresponds to the voltage measurements taken atlocation 124 (the “gs_comp” signal line in the pad driver 34) in FIG. 9,the voltage plot 125 refers to the Vpad measurement at the circuit pad“padd” 125 (similar to the IC pad 24 in FIG. 3), and so on. In the samemanner, the current diagram 134 includes a number of plots (except forthe plots 91, 93, and 95) bearing reference numerals corresponding tothe locations of current measurements in FIG. 9. For example, thecurrent graph 125 corresponds to the current measurements taken at the“padd” terminal 125 in FIG. 9, thereby taking the Rpad into account.Similarly, the current plot 128 corresponds to the current measurementstaken at the “padd-unc” terminal 128 in FIG. 9, thereby taking the Rpadof the uncompensated driver 72 into account. The ideal plots 91, 93, and95 are already noted hereinbefore with reference to FIG. 5B. It is seenfrom the voltage plot 123 in FIG. 10A for the slow SS corner (worst forMOS drive capability) that the Vgs of the pad driver NMOS (i.e.,transistor M102) is well below supply Vdd (which is depicted at graph 83in FIG. 5A), at just 1.6V. Even so, curvature compensation is not reallyneeded for the Vpad voltage range of interest (i.e., below the 22 mAline 95 in the diagram 134). Because of the ‘double amplifier offset’(resulting from the use of the primary amplifier 60 and the biasamplifier 118) discussed hereinbefore, both the compensated (plot 125)and uncompensated (plot 128) pad impedances are not well centered at thepoint of calibration, which is Vpad=Vref=1.1V.

FIGS. 11A and 11B illustrate two diagrams 136, 138 showing plots ofvoltages and currents (versus time), respectively, at various terminalsor points in the MOS driver circuit 114 in FIG. 9 when the circuit 114is SPICE simulated in the fast (FF) corner, with rising edge for Vpad.The voltages in diagram 136 are measured in volts and mV, whereas thecurrents in diagram 138 are measured in mA. The time is given innanoseconds in both diagrams. Various plots or graphs in the diagrams136, 138 are referenced in the same manner as that described above withreference to FIGS. 10A-10B. Therefore, no additional explanation ofvarious plots in FIGS. 11A-11B is provided. As can be seen from plot 123in FIG. 11A, in the fast FF corner (best for MOS drive capability), theVgs of the pad driver NMOS (i.e., transistor M102) has dropped to just abit more than 1.0V. This leads to a large linear region MOS impedancecurvature for the uncompensated pad (as can be seen from graph 128 inFIG. 11B). The compensated pad (i.e., the circuit 114), however, islinear as can be seen from graph 125 in FIG. 11B. This illustrates theeffect of curvature compensation according to the present disclosure.When it is difficult to drive a MOS gate voltage to full Vdd supply(perhaps because of reliability concerns with technology limits, orbecause of the use of a very thin oxide high drive MOS (to a terminatingvoltage<Vdd) for maximum drive), then using the methodology described inthe present disclosure it may be possible to eliminate the MOS linearregion curvature that would otherwise make the impedance spread toogreat.

Third Circuit Realization of FIG. 3

FIG. 12 is an exemplary schematic of a practical realization of the MOSdriver circuit 32 in FIG. 3 according to one embodiment of the presentdisclosure. The “compensated” pad driver circuit (or MOS driver circuit)140 in FIG. 12 is shown along with a conventional “uncompensated” paddriver 72 (similar to the driver circuit 12 in FIG. 2) just forcomparison purpose. As FIG. 12 is a realization of the general circuitconfiguration in FIG. 3, the circuit blocks having similar functions inFIGS. 3 and 12 are denoted by the same reference numerals withoutassigning reference numerals to each circuit component in FIG. 12 (e.g.,resistor, transistor, capacitor, etc.). Further, as the function of thecircuit in FIG. 12 can be easily explained when a comparison of FIGS. 3and 12 is made, a detailed description of the circuit components in FIG.12 is omitted here for the sake of brevity. From a comparison of FIGS. 3and 12, it is seen that the MOS driver circuit 140 in FIG. 12 includesthe pad driver 34, the replica of the pad driver 36, the voltage summercircuit 46 (implemented in a voltage follower configuration), theamplifier 60, and an additional amplifier 142 having a referencefollower circuit 141. It is noted that the amplifier 60 in theembodiment of FIG. 12 is a two-stage amplifier, whereas the sameamplifier is shown as a single stage amplifier in the embodiment of FIG.9. The amplifier 142 is an additional two-stage amplifier connectedbetween the replica pad driver 36 and the combination of the pad driver34 and the voltage summer 46 to provide a bias for the follower circuit46 (on the line indicated by reference numeral “152” in FIG. 12) and theVgs (on line 144) for the pad driver NMOS transistor M102. Thus, theinverting input 47 (FIG. 3) to the voltage summer 46 is not provideddirectly from the pad driver replica 36, but through the amplifier 142.It is observed that the Vref_fb signal is provided to the amplifier 60when the test pads 146 and 147 are connected. Despite someimplementational variations, the driver circuit 140 in FIG. 12 isfunctionally equivalent to the circuit 32 in FIG. 3.

The MOS driver circuit 140 in FIG. 12 is considerably more complex thanthe realization shown in FIG. 4. The MOS driver circuit 140 of FIG. 12may be used, with its PMOS input stage (in amplifier 60), when a lowreference voltage (e.g., Vref=0.5V) is available. However, in thesimulation of circuit 140, the value of 1.1V was used for Vref. Somepoints to note about the MOS driver 140 in FIG. 12 are: (1) Unlike theprevious circuit 114 of FIG. 9, the circuit 140 in FIG. 12 may be ableto drive the gate of the NMOS (line 144 of transistor M102) close to Vddbecause of the extra amplifier (level shifting) stage (i.e., the secondstage in the two-stage amplifier 142). (2) Additional stabilization maybe desirable for the circuit 140 in the FF corner. (3) The increasedvoltage gain of the two-stage amplifier 60 may minimize input offsets.Therefore, it may be possible to use a second amplifier (here, theadditional two-stage amplifier 142) to generate the follower bias (online 152), without the compound amplifier offsets being too high. (4)The biased two-stage amplifier 142 may be a more robust solution thanthe unbiased amplifier 118 used in FIG. 9. (5) It may be desirable toconnect the output of the main amplifier 60 to the gate of the paddriver 34 NMOS transistor M102 (i.e., the ‘ngate’ node 144 in FIG. 12)and perform a pole-zero cancellation technique for compensation (R-Crather than simple C), instead of using the second two-stage amplifier142 to supply the Vgs for pad driver NMOS. Alternatively, it may bepossible to use a two stage push-pull amplifier (not shown) (first stagediode connected, and the second stage cascoded for high outputresistance) for the amplifier circuit 60 and drive the heavily loaded‘ngate’ node 144 directly (i.e., without using the amplifier 142 tosupply the Vgs for pad driver NMOS transistor M102).

SPICE Simulation Results for the Circuit 140 of FIG. 12

FIGS. 13A-13C illustrate three diagrams 160, 162, 164 showing plots ofvoltages and currents (versus time) at various terminals or points inthe MOS driver circuit 140 in FIG. 12 when the circuit 140 is SPICEsimulated in the slow SS corner, with rising edge for Vpad. The voltagesin diagrams 160, 162 are measured in volts and mV, whereas the currentsin diagram 164 are measured in mA and μA. The time is given innanoseconds in all diagrams. Various plots or graphs in the voltagediagrams 160, 162 in FIGS. 13A-13B are referenced by the same numeralsas those used to identify corresponding test locations in the circuit inFIG. 12 where the voltage measurements are taken. For example, thevoltage plot 153 corresponds to the voltage measurements taken atlocation 153 (the “ds_comp” signal line in the pad driver 34) in FIG.12. Similarly, the voltage plot 149 corresponds to the voltagemeasurements taken at location 149 (the “stg1” signal line in theamplifier 60) in FIG. 12, the voltage plot 151 refers to the voltagemeasurement at the “ngate_rep” terminal 151 in the pad driver circuit34, and so on. In the same manner, the current diagram 164 in FIG. 13Cincludes a number of plots (except for the plots 91, 93, and 95) bearingreference numerals corresponding to the locations of currentmeasurements in FIG. 12. For example, the current graph 155 correspondsto the current measurements taken at the “padd” terminal 155 in FIG. 12(similar to the IC pad 24 in FIG. 3), thereby taking the Rpad intoaccount. Similarly, the current plot 156 corresponds to the currentmeasurements taken at the “padd-unc” terminal 156 in FIG. 12, therebytaking the Rpad of the uncompensated driver 72 into account. The idealplots 91, 93, and 95 are already noted hereinbefore with reference toFIG. 5B.

FIGS. 14A-14C illustrate three diagrams 166, 168, 170 showing plots ofvoltages and currents (versus time) at various terminals or points inthe MOS driver circuit 140 in FIG. 12 when the circuit 140 is SPICEsimulated in the fast (FF) corner, with rising edge for Vpad. Thevoltages in diagrams 166, 168 are measured in volts and mV, whereas thecurrents in diagram 170 are measured in mA and μA. The time is given innanoseconds in both diagrams. Various plots or graphs in the diagrams166, 168, 170 are referenced in the same manner as that described abovewith reference to FIGS. 13A-13C. Therefore, no additional explanation ofvarious plots in FIGS. 14A-14C is provided.

It is observed that various current and voltage graphs in FIGS. 13A-13C,14A-14C are substantially similar to various graphs in FIGS. 5A-5B,6A-6B representing voltages and currents at corresponding locations inthe circuit 68 in FIG. 4. Because of the similarity of FIGS. 13A-13C,14A-14C and FIGS. 5A-5B, 6A-6B, no additional explanation is providedfor plots in FIGS. 13A-13C, 14A-14C. Thus, although the circuit 140 inFIG. 12 is considerably more complex than the circuit 68 in FIG. 4, theoverall performance of both of those circuits is substantially similar.

Performance of Circuit 68 in FIG. 4 in HS mode

It is noted that in USB2.0, for example, not only does pad impedance(Rpad) have to be controlled to within 45 Ohms ±10% during FS (FullSpeed, 12 Mega Bits per second) mode of operation, but that the MOSdriver/buffer (e.g., the circuit 32 in FIG. 3) in the FS mode may alsobe used as a ground reference impedance for HS (High Speed, 480 MegaBits per second) mode of operation of the MOS buffer.

FIGS. 15A-15B illustrate slow (SS) corner results of SPICE simulation ofthe MOS driver circuit 68 in FIG. 4 in the HS mode, whereas FIGS.16A-16B illustrate fast (FF) corner results of SPICE simulation of theMOS driver 68 in FIG. 4 in the HS mode. FIGS. 15A-15B and 16A-16B showdiagrams 172, 174, 176, 178 of voltages and currents (versus time) atvarious terminals or points in the MOS driver circuit 68 in FIG. 4 whenthe circuit 68 is SPICE simulated in the HS mode. The voltages indiagram 172 are measured in volts, whereas they are measured in V and mVin diagram 176. The currents in diagram 174 are measured in amperes,whereas they are measured in μA and mA in diagram 178. The time is givenin seconds in diagrams 172, 174, whereas it is given in nanoseconds indiagrams 176, 178. Various plots or graphs in the voltage diagrams 172,176 are referenced by the same numerals as those used to identifycorresponding test locations in the circuit in FIG. 4 where the voltagemeasurements are taken. For example, the voltage plot 90 corresponds tothe voltage measurements (Vpad) taken at location “padd” 90 (similar tothe IC pad 24 in FIG. 3) in FIG. 4. Similarly, the voltage plot 87corresponds to the voltage measurements taken at location 87 (the“fb_int_comp” line) in FIG. 4, the voltage plot 76 corresponds to thevoltage measurements taken at location 76 (the “y” line) in FIG. 4, andso on. In the same manner, the current diagrams 174, 178 include plots90 that correspond to the measurements of current flowing through theinternal linearizing resistor Rs 22 (FIG. 4), i.e., current measurementstaken at the “padd” terminal 90 in FIG. 4, thereby taking the Rpad intoaccount.

The results depicted in FIGS. 15A-15B and 16A-16B were obtained byinstantiating two parallel USB1.1 buffers (not shown) (each buffersimilar to the circuit 68 in FIG. 4) with combined nominal resistance of22.5 Ohms. A current source of 17.77 mA was injected into the buffers at480 MHz, with rise and fall times of 500 ps. The slow and fast cornerresults of the simulation are shown in FIGS. 15A-15B and 16A-16Brespectively. Both results show that the output high level of impedanceRpad is well within the USB2.0 specification limits of 45 Ohms ±10%.However, in all of FIGS. 15A-15B and 16A-16B the levels of impedancevalues are slightly low. This may be corrected by circuit optimizations,for example, by changing Vref and/or changing the values of seriesresistors Rs and 2*m*Rs (shown in FIG. 3). The waveforms in FIGS.15A-15B and 16A-16B are for a scenario where an impedance controlled paddriver circuit according to the present disclosure (e.g., the parallelUSB1.1 buffers mentioned at the beginning of this paragraph, with 45Ohms ground referenced resistive termination for the HS mode) is used asa termination resistor. The simulations in FIGS. 15A-15B and 16A-16Bemulate HS mode data transmission in USB2.0, with a 480 MHz clocksignal.

It is observed that it may be desirable to build a “high side” impedancecontrol circuit for a MOS driver circuit built according to theteachings of the present disclosure (i.e., built similar to the drivercircuit 32 in FIG. 3). A “high side” impedance control circuit may beused to control the high driving impedance of an IC pad. Such animpedance control circuit may be designed as a mirror image of the MOSdriver circuit in FIG. 3. As is known in the art, in the mirror circuit(mirrored in the y-axis) (not shown), for example, transistors M1, Maand Mb in FIG. 3 would become PMOS devices, the current source 58 wouldchange to a current sink, etc. For example, FIG. 17 (discussed below)shows how to generate the reference voltages (V_(REF) _(—) _(L) andV_(REF) _(—) _(H)) for low and high side impedance control.

It is further observed that throughout the discussion of various MOSdriver circuit configurations provided herein with reference to FIGS. 3,4, 9, and 12, the nominal resistor values (for various resistors in thecircuits described hereinbefore) were chosen at 0° C. In case of Rpadunder USB2.0 specification, for example, this may mean that a ±10%process spread and a −5% (−40° C.) to +20% (105° C.) temperature spread,the total effective spread may be −15% +30%. Therefore, for a resistanceRpad centered on 45 Ohms (as required under USB2.0, for example), it maybe preferable to choose a nominal resistance value of about 42 Ohms toaccommodate these PT spreads and still comply with the USB2.0requirements. This nominal resistance value correction factor may beapplied to the circuits in FIGS. 3, 4, 9, and 12.

Probable Bandgap Reference Considerations

It is noted that in the exemplary MOS driver circuits in FIGS. 4, 9, and12, a 1.1V reference voltage Vref has been used. However, some of thecircuits (e.g., circuit 114 in FIG. 9 and circuit 140 in FIG. 12) may bemade to work with a lower reference Vref (for example, 0.55V). Thesimplest circuit 68 of FIG. 4 may not function with a reference voltageVref much below 1.1 V, because of its NMOS input stage (in amplifier70). It is observed, however, that a higher voltage reference (e.g.,Vref=Vdd−1.1V) may be needed for the high side impedance Control notedhereinbefore. Therefore, it may not be necessary to choose a referencebest suited for impedance control, because the bandgap may have toproduce >1 voltage references.

FIG. 17 illustrates how, when a bandgap circuit operates on a supplyother than the 3.3V used for a pad driver circuit according to thepresent disclosure (e.g., on a 1.5V supply), one can effectively “levelshift” the bandgap voltage to generate the required high and low valuesfor the reference voltage (Vref) to a MOS pad driver circuit (e.g., anyMOS driver circuit based on the general circuit configuration 32 in FIG.3) constructed according to teachings of the present disclosure. Theexemplary circuit configuration 180 shown in FIG. 17 includes anadditional stage 182 to the bandgap circuit (not shown) running on a1.5V analog supply voltage and a cascode stage 184 (for currentprecision) in series with the additional stage 182. The bandgap circuit(not shown) precedes the additional stage 182. The resistors RBG, R1,and R2 are matched and inter-digitated (i.e., laid out together, ratherthan individually). The circuit 180 is shown to provide two referenceoutputs: the low reference voltage Vref_L=1.1V, and the high referencevoltage Vref_H=Vdd−1.1V. The Vref_L voltage may be supplied as a Vrefinput to a MOS driver circuit constructed according to the teachings ofthe present disclosure (e.g., the Vref input 62 in the circuit 32 inFIG. 3, or the “ref” reference voltage terminal 85 in the circuit 68 inFIG. 4, etc.), whereas the Vref_H voltage may be fed as a referencevoltage to a y-axis mirror image (not shown) of the MOS driver circuit(e.g., the circuit 32 in FIG. 3) discussed hereinbefore with referenceto high side impedance control.

Using USB1.1 Buffer Stand Alone (not within a USB2.0 PHY)

A MOS driver circuit constructed according to the teachings of thepresent disclosure (e.g., the MOS driver circuit 32 in FIG. 3) may beused as a stand alone USB1.1 buffer/driver, instead of a USB1.1 bufferwithin a USB2.2 PHY device. Two exemplary ways of using the drivercircuit 32 as a stand alone USB1.1 buffer are discussed hereinbelow withreference to FIGS. 18 and 19.

FIG. 18 illustrates the MOS driver circuit 32 in FIG. 3 with the currentreference “Iref” supplied by a separate bias generator 186 so as toallow the use of the driver circuit 32 as a stand alone USB1.1 buffer.As the MOS driver circuit 32 in FIG. 18 is the same circuit shown inFIG. 3, only some of the major circuit elements are identified withreference numerals for the sake of clarity. The separate bias generator186 may include an external precision resistor 188, an NMOS transistor192, a differential amplifier 190, and a voltage divider networkconsisting of resistors R1 (194) and R2 (196). An internal potentialdivider (of resistors R1 and R2) may allow Vref to be stable within±10%. The Vref generated at the junction of resistors R1 (194) and R2(196) may be supplied to the non-inverting terminal 62 of the op-amp 60,and the current reference Iref 58 may be obtained from the currentflowing into the drain of transistor 192.

The separate bias generator 186 may be available in the market as aUSB1.1 pad. The bias pad 186 may be a dedicated pad with the externalprecision (within 1%) resistor 188. The bias pad 186 need not be abandgap reference generator. In one embodiment, the bias generator 186may have five (5) current outputs (not shown) for up to five USB1.1 MOSdrivers (each similar to the driver 32). In the arrangement shown inFIG. 18, the value of internal linearizing resistor Rs may have to bereduced (from 28 to 22 Ohms) to meet the lower overall USB1.1 standalone impedance specification of 28 to 44 Ohms. It is noted that thisnew Rs value of 22 Ohms may apply to a common schematic for thestand-alone version of the MOS driver circuit 32 (e.g., theconfiguration illustrated in FIG. 18) and a version of the circuit 32used within a USB2.0 PHY (e.g., the configurations illustrated in FIGS.4, 9, and 12). This may however mean that the gate of a MOS pad driver(e.g., the gate 39 of transistor Ma in FIG. 3) may be driven slightlylower than the optimal maximum (Vgs). That may be of no concern,however, as the curvature adjust mechanism may not be affected by slightdeviations from the maximum Vgs.

Some points to note about the separate bias generator configuration inFIG. 18 are: (1) No external termination resistors (similar to theinternal linearizing resistor Rs) may be required for DP and DM IC pads.It is observed here that in USB1.1 and USB2.0, differential IC pads areused, i.e., there are actually two IC pads (DP or D+ and DM or D−), eachpad connected to a MOS driver circuit like the circuit 32 in FIG. 3.Therefore, for example, in case of five (5) USB1.1 buffers (each similarto circuit 32), it may be possible to save ten (10) such externalresistors because, as noted before, a single bias generator 186 mayprovide reference currents to all five USB1.1 buffers. (2) The same biasgenerator pad 186 may be used for a USB1.1 stand alone buffer and for aUSB1.1 buffer inside a USB2.0 PHY. (3) Reasonably precise control ofslew rates may be obtained from relatively precise current referenceprovided by the bias pad 186. (4) The approach illustrated in FIG. 18requires use of an additional pad 198 to which an external precisionresistor 188 must be connected. However, as noted before, only oneadditional pad may be needed for up to five USB1.1 buffers. (5) The useof a separate bias pad 186 for current reference may require a slightlylarger (10%) MOS transistor M1 (52) in the pad driver replica 36. Unlikewhen Vref is generated by a band-gap (e.g., as discussed hereinbeforewith reference to FIG. 17), in the scheme of FIG. 18, Vref has a ±10%spread because it is generated using a simple potential divider of R1(194) and R2 (196). Therefore, the voltage at the replica IC pad 198will vary from approximately 1.2V to 1.1V to 1.0V as Vdd varies from3.6V to 3.3V to 3.0V. The voltage across the external precision resistor188 will therefore vary by ±10% (with nominal value obtained when Vdd isat its nominal value of 3.3V) and so the current through the externalresistor 188 will vary by ±10%. Therefore, Iref 58 varies by ±10%. Ifthe reference current (Iref 58) is 10% higher than the nominal, then thegate voltage of M1 (52) must increase to pass the increased current. IfVgs of M1 cannot however increase (e.g., because the amplifier 60 cannotoutput more voltage), then the size of the M1 must be increased by 10%.

In the USB1.1 stand alone operation depicted in the embodiment of FIG.18, it may be desirable to choose a higher nominal Iref value so thatpad impedance (Rpad) at IC pad 24 will be less than 44 Ohms (the upperlimit on the stand alone overall impedance specification under USB1.1).For example, suppose that 2*m=24 (m=12 fingers) and that nominal Vref(Vref.nom) =1.1V. (1) In case of a USB1.1 buffer within a USB2.0 PHY,replica pad resistance=2*m*Rs=24 * 22=528 Ohms. Here, the impedance ofNMOS M1 (52) may be adjusted such that it provides 552 Ohms, making atotal of 1080 Ohms pad driver replica resistance. Hence, Iref may bechosen equal to 1.02 mA so that Vref.nom=Vref_fb=1.02 mA * 1080Ohms=1.1. (2) On the other hand, for USB1.1 buffer in stand aloneconfiguration (similar to that shown in FIG. 18), the nominal value ofIref may be chosen, for example, 15% higher at 1.15 mA. The replica padresistance remains the same at 2*m*Rs=24 * 22=528 Ohms and the impedanceof NMOS M1 (52) may be adjusted such that it provides 428 Ohms, therebymaking the total replica resistance of 956 Ohms. This means thatVref.nom=Vref_fb=1.15 mA*956=1.1V, which is suitable for the stableoperation of the amplifier 60. With these values, Rpad turns out to beequal to 956/24=40 Ohms, which is well below the 44 Ohms upper limit(noted before) for a USB1.1 stand alone buffer. Because the percentagechange in Iref (±10%) is the same as that in Vref, overall trimmedimpedance (Rpad) may remain constant with changes in Vdd. Hence, nobandgap voltage may be needed for the bias generator 186.

FIG. 19 illustrates a MOS driver circuit 200, which is identical to MOSdriver 32 in FIG. 3, but with its impedance control inhibited so as toallow the use of the driver circuit 32 as a stand alone USB1.1 buffer.In the embodiment of FIG. 19, the MOS driver 32 of FIG. 3 is modified sothat no amplifiers shown in FIG. 3 (e.g., amplifiers 46, 60) are active(thereby inhibiting impedance control in the circuit 32 of FIG. 3). Ascan be seen from FIG. 19, the gates 39, 43 of the pad driver NMOStransistors Ma (38) and Mb (42), respectively, are directly driven toVdd. In the embodiment of FIG. 19, an external 22 Ohm precision resistor206 is used (in practice, two such 22 Ω external resistors—one for DPand one for DM—may be used). A portion 204 of the internal linearizingresistor Rs 22 is deactivated or shorted out (e.g., using a metalstrap). As the circuit 32 in FIG. 3 may function as a USB1.1 bufferwithin a USB2.0 PHY device (as discussed before), the internallinearizing resistor Rs 22 may be equal to 28 Ohms. Therefore, theportion 204 of the internal resistor Rs shorted out may equal to 20Ohms, and the other (active) portion 202 may equal 8 Ohms for ESD(Electro Static Discharge) of any sporadic large currents. The otherportion 202 of the internal resistor Rs 22 is therefore used in serieswith the external precision resistor 200 to obtain the desired overallpad impedance (Rpad). For example, the combined resistance of NMOStransistors Ma 38 and Mb 42 may be less than 9 Ohms, whereas theimpedance spread of the active internal linearizing resistor 202 may befrom 7 Ohms (minimum) to 10 Ohms (maximum). In that event, the overallimpedance spread for Rpad (taking into account the external seriesresistor of 22 Ohms) will be from about 38 to 41 Ohms, which is withinthe USB1.1 stand alone impedance specification of 28 to 44 Ohms.

Some points to note about the circuit configuration of FIG. 19 are: (1)There is no need for a separate bias generator (as, for example, in theconfiguration of FIG. 18), thereby saving one pad on the chip design.(2) The control of slew rates may be less precise because no precisecurrent reference is available. (3) Because of the use of externaltermination resistors 206 for DP and DM, there may be a need for 10external resistors for 5 USB1.1 buffers in the configuration of FIG. 19.(4) Because of the metal strap shorting part of the internal lineraizingresistor Rs 22, the circuit layouts during chip fabrication may bedifferent for the USB1.1 stand alone buffer in FIG. 19 and for a USB1.1buffer (e.g., the circuit 32 in FIG. 3) inside a USB2.0 PHY.

It is noted here that all the circuit configurations shown and discussedhereinabove are as examples only. Therefore, although all the resistorsin various figures in the present disclosure are shown as passiveresistors, it is known to one skilled in the art that appropriateresistance may be obtained by fabricating a resistor using one or moreactive components (e.g., diodes, transistors, etc.). It is also pointedout that various NMOS transistors shown in different figures in thepresent disclosure may be replaced with suitably biased PMOS transistorsand vice versa, as is known in the art. Further, except for the externalcomponents needed to complete an operational circuit (e.g., the externalbias pad 186 in the embodiment of FIG. 18, or the external precisionresistor 206 in the embodiment of FIG. 19, etc.), all of the circuitconfigurations shown in various figures (e.g., FIG. 3, FIG. 4, FIG. 9,etc.) in the present disclosure may be fabricated on a single siliconchip as is also known in the art. It is further noted that although thediscussion herein of MOS linear region impedance curvature controlmethods has been with reference to the pad impedance specification underUSB2.0, it will be apparent to one skilled in the art to employ thepresent impedance control methods and various circuit approaches (withsuitable modifications known in the art) to obtain MOS driver/buffercircuits complying with any other specific requirements (which may bedifferent from the requirements under USB2.0). It is still further notedthat all SPICE simulations discussed throughout the disclosure wereperformed using a 0.11 μm CMOS fabrication process.

FIG. 20 is a block diagram depicting a system 220 in which a MOS drivercircuit constructed according to the teachings of the present disclosuremay be used. The system 220 may include a data processing unit orcomputing unit 222 that includes a processor 224 for performing variouscomputing functions, such as executing specific software to performspecific calculations or data processing tasks. The computing unit 222also includes a memory controller 226 that is in communication with theprocessor 224 through a bus 230. The bus 230 may include an address bus(not shown), a data bus (not shown), and a control bus (not shown). Thememory controller 226 is also in communication with a set of memorydevices 228 through another bus 232. In one embodiment, each memorydevice 228 is a synchronous dynamic random access memory (SDRAM). Eachmemory device 228 may include appropriate data storage and retrievalcircuitry (not shown) as is known in the art. The processor 224 canperform a plurality of functions based on information and data stored inthe SDRAMs 228. The system 220 may include one or more input devices 234(e.g., a keyboard or a mouse) connected to the computing unit 222 toallow a user to manually input data, instructions, etc., to operate thecomputing unit 222. One or more output devices 238 connected to thecomputing unit 222 may also be provided as part of the system 220 todisplay or otherwise output data generated by the processor 224.Examples of output devices 238 include printers, video terminals orvideo display units (VDUs). In one embodiment, the system 220 alsoincludes one or more data storage devices 236 connected to the dataprocessing unit 222 to allow the processor 224 to store data in orretrieve data from internal or external storage media (not shown).Examples of typical data storage devices 236 include drives that accepthard and floppy disks, CD-ROMs (compact disk read-only memories), andtape cassettes. An output buffer or driver circuit in any of the circuitelements in system 220—the processor 224, the memory controller 226, oneor more memory devices 228, one or more input devices 234, one or moreoutput devices 238, and one or more storage devices 236—may beconfigured to include the MOS driver circuit (e.g., the circuit 32 inFIG. 3 or any of its implementations in FIG. 4, 9 or 12) according tothe present disclosure. Some examples of the devices that may includethe MOS driver circuit (e.g., the driver circuit 32 in FIG. 3) aredevices such as printers, computer mouse, webcams, camcorders, etc.,which use USB1.1 or USB2.0 IC's.

The foregoing describes a system and method to correct or cancel MOSlinear region impedance curvature. An analog solution (given its manyadvantages in terms of, for example, reduced complexity and conservationof silicon area) is provided to trim out the MOS linear region impedancecurvature while accommodating PVT spreads in values of internal orexternal precision resistors. The linear region curvature correction maybe obtained by using two MOS transistors in the pad driver/buffer andoperating the transistors so as to proportionately increase outputimpedance of one of them when the output impedance of the otherdecreases, and vice versa. A linear pad impedance may be maintained overa range of Vpad values, while also maintaining the Vgs supplied to paddriver transistors at its maximum possible value to obtain greaterlinearity. The approach of the present disclosure relaxes therequirements on the voltage/current references used in the MOS paddrivers and makes tight impedance control possible, especially in asituation where the MOS fabrication process (typically all currentlyused processes) does not have available an internal precision resistorwith a reasonably well controlled value.

While the disclosure has been described in detail and with reference tospecific embodiments thereof, it will be apparent to one skilled in theart that various changes and modifications can be made therein withoutdeparting from the spirit and scope of the embodiments. Thus, it isintended that the present disclosure cover the modifications andvariations of this disclosure provided they come within the scope of theappended claims and their equivalents.

1. A MOS driver circuit comprising: a first MOS transistor configured toreceive a controlled voltage at a first input terminal thereof; and asecond MOS transistor coupled to said first MOS transistor andconfigured to receive a differential voltage at a second input terminalthereof, wherein said second MOS transistor is configured to have anincreased output impedance when output impedance of said first MOStransistor decreases, and vice versa; a signal adder circuit coupled tosaid first and said second MOS transistors, wherein a first output ofsaid signal adder circuit is coupled to said second input terminal toprovide said differential voltage to said second MOS transistor, andwherein a first output terminal of said first MOS transistor is coupledto a first input of said signal adder circuit to provide a first inputvoltage thereto; a scaled replica of said MOS driver circuit having asecond output coupled to a second input of said signal adder circuit toprovide a second input voltage thereto; and a linearizing resistorcoupled in series with said first output terminal of said first MOStransistor and a second output terminal of said second MOS transistor.2. The MOS driver circuit of claim 1, wherein said scaled replicaincludes a third MOS transistor, wherein a third output terminal of saidthird MOS transistor is configured to function as said second output. 3.The MOS driver circuit of claim 2, wherein said first, said second, andsaid third MOS transistors are CMOS transistors.
 4. The MOS drivercircuit of claim 2, wherein said third output terminal is a drainterminal of said third MOS transistor.
 5. The MOS driver circuit ofclaim 2, wherein source terminals of said first, said second, and saidthird MOS transistors are held at a common reference potential.
 6. TheMOS driver circuit of claim 2, further comprising a current referencecoupled to said third output terminal of said third MOS transistor. 7.The MOS driver circuit of claim 2, further comprising: a first amplifierhaving a third output coupled to a third input terminal of said thirdMOS transistor to supply said controlled voltage to said third inputterminal; and a second amplifier coupled to said third MOS transistorand said first MOS transistor, wherein said second amplifier isconfigured to provide said controlled voltage to said first inputterminal of said first MOS transistor.
 8. A MOS driver circuitcomprising: a first MOS transistor configured to receive a controlledvoltage at a first input terminal thereof; a second MOS transistorcoupled to said first MOS transistor and configured to receive adifferential voltage at a second input terminal thereof, wherein saidsecond MOS transistor is configured to compensate for changes in outputimpedance of said first MOS transistor through corresponding changes inoutput impedance of said second MOS transistor; a signal adder circuitcoupled to said first and said second MOS transistors, wherein a firstoutput of said signal adder circuit is coupled to said second inputterminal to provide said differential voltage to said second MOStransistor, and wherein a first output terminal of said first MOStransistor is coupled to a first input of said signal adder circuit toprovide a first input voltage thereto; a scaled replica of said MOSdriver circuit having a second output coupled to a second input of saidsignal adder circuit to provide a second input voltage thereto; and alinearizing resistor coupled in series with said first output terminalof said first MOS transistor and a second output terminal of said secondMOS transistor.
 9. The MOS driver circuit of claim 8, wherein saidscaled replica includes a third MOS transistor, wherein a third outputterminal of said third MOS transistor is configured to function as saidsecond output.
 10. The MOS driver circuit of claim 9, furthercomprising: a first amplifier having a third output coupled to a thirdinput terminal of said third MOS transistor and said first inputterminal of said first MOS transistor to supply said controlled voltageto said first and said third input terminals.
 11. The MOS driver circuitof claim 10, wherein said first amplifier is an inverting differentialamplifier.
 12. The MOS driver circuit of claim 10, wherein said firstamplifier has at least two inputs, and wherein at least one of said atleast two inputs is held at a reference potential.
 13. The MOS drivercircuit of claim 10, wherein an input of said first amplifier and saidthird output terminal of said third MOS transistor are coupled to a DCsupply voltage.
 14. The MOS driver circuit of claim 10, furthercomprising: a second amplifier connected between said second output ofsaid scaled replica and said second input of said signal adder circuit,wherein said second amplifier is configured to provide said second inputvoltage to said signal adder circuit.
 15. A method of operating a MOSdriver circuit, said method comprising: using a first MOS transistorwith a first terminal, a second terminal, and a third terminal; using asecond MOS transistor with a fourth terminal, a fifth terminal, and asixth terminal; providing a DC supply voltage to said first and saidfourth terminals; further providing a reference potential to said secondand said fifth terminals; and further using an internal linearizingresistor and an external precision resistor in series with said thirdand said sixth terminals.
 16. The method of claim 15, wherein saidfurther using includes: deactivating a portion of said internallinearizing resistor during operation of said MOS driver circuit. 17.The method of claim 15, wherein said further using includes using only aportion of said internal linearizing resistor in series with said thirdand said sixth terminals.
 18. The method of claim 17, wherein saidportion of said internal linearizing resistor has a nominal value of 8Ohms, and wherein said external precision resistor has a nominal valueof 22 Ohms.